資源目錄里展示的全都有，所見即所得。下載后全都有,請放心下載。原稿可自行編輯修改=【QQ：401339828 或11970985 有疑問可加】 任務(wù)書 一、畢業(yè)設(shè)計(jì)（論文）的內(nèi)容 本系統(tǒng)在進(jìn)行實(shí)際調(diào)研和廣泛收集相關(guān)資料的基礎(chǔ)上，設(shè)計(jì)C650立式車床電氣控制系統(tǒng)（包括主電路及其控制系統(tǒng)），主電機(jī)要求采用Y-D起動控制；設(shè)計(jì)短路保護(hù)、過流保護(hù)、欠壓保護(hù)等，計(jì)算、分析并選擇電動機(jī)、斷路器、交流接觸器、繼電器、互感器、電纜線等主要電器的電流、功率、型號、等級等具體數(shù)據(jù)，編制C650立式車床電氣控制系統(tǒng)PLC程序清單；設(shè)計(jì)并制作C650立式車床電氣控制模擬樣機(jī)，并采用PLC編程實(shí)現(xiàn)C650立式車床的各種控制。 二、畢業(yè)設(shè)計(jì)（論文）的要求與數(shù)據(jù) 設(shè)計(jì)C650立式車床電氣控制系統(tǒng)（包括主電路及其控制系統(tǒng)），主電機(jī)要求采用Y-D起動控制；計(jì)算、分析并選擇電動機(jī)、斷路器、交流接觸器、繼電器、互感器、電纜線等主要電器的電流。功率、型號、等級等具體數(shù)據(jù)，編制C650立式車床電氣控制系統(tǒng)PLC程序清單；要求設(shè)計(jì)過流保護(hù)：運(yùn)行中當(dāng)電機(jī)電流大于額定電流的1.1倍且持續(xù)時間達(dá)1分鐘則產(chǎn)生過流跳閘,同時產(chǎn)生聲光報(bào)警信號；設(shè)計(jì)主電機(jī)短路保護(hù)；設(shè)計(jì)電源電壓欠壓保護(hù)：當(dāng)電源電壓小于額定電壓的0.75倍時，電氣控制系統(tǒng)產(chǎn)生欠壓保護(hù)，同時發(fā)出相應(yīng)的聲光報(bào)警信號；要求設(shè)計(jì)制作C650立式車床電氣控制系統(tǒng)模擬樣機(jī)，編制C650立式車床電氣控制系統(tǒng)PLC程序清單。 三、畢業(yè)設(shè)計(jì)（論文）應(yīng)完成的工作 1.完成四萬印刷字符的英文資料翻譯（要求打印，并要求附英文原文）； 2.撰寫兩萬字左右的畢業(yè)設(shè)計(jì)論文（兼附15篇以上的參考文獻(xiàn)；論文要求打?。?； 3.在畢業(yè)設(shè)計(jì)論文中必須包括300-500個單詞的英文摘要； 4.設(shè)計(jì)C650立式車床電氣主回路系統(tǒng)； 5.設(shè)計(jì)C650立式車床電氣控制系統(tǒng)； 6.編制C650立式車床電氣控制系統(tǒng)PLC程序清單； 7.用計(jì)算機(jī)繪制3#圖紙至少2張； 8.設(shè)計(jì)制作C650立式車床電氣控制系統(tǒng)模擬樣機(jī)。 四、應(yīng)收集的資料及主要參考文獻(xiàn) [1]楊林建.機(jī)床電氣控制技術(shù)[M].北京市：北京理工大學(xué)出版社?,2008? [2]盛炳乾,李軍.工業(yè)過程測量與控制[M].北京：中國輕工業(yè)出版社，1996 [3]楊興瑤.電氣傳動及應(yīng)用[M].北京：化學(xué)工業(yè)出版社，1994. [4]顧繩谷.電機(jī)及拖動基礎(chǔ)（下）[M].合肥：合肥工業(yè)大學(xué)，2000. [5]陳隆昌,閻治安,劉新正.控制電機(jī)[M].西安：西安電子科技大學(xué)出版社，2000 [6]齊從謙，王士蘭. PLC技術(shù)及應(yīng)用[M].北京：機(jī)械工業(yè)出版社，2000.8 [7]程周，毛臣健，章小印. 電氣控制與PLC原理及應(yīng)用[M].北京：電子工業(yè)出版社，2003 [8]李乃夫.可編程控制器原理、應(yīng)用、實(shí)驗(yàn)[M].北京：中國輕工業(yè)出版社,2003.2 [9]鄒金慧,黃宋魏,楊曉洪.可編程序控制器（PLC）原理及應(yīng)用[M].北京：機(jī)械工業(yè)出版社,2001 [10] Mohamed EA Rand. Artificial neural network based faulted diagnostic system for electric power distribution feeders [J].Electric Power Systems Research, 1995,35(4):1—10 五、試驗(yàn)、測試、試制加工所需主要儀器設(shè)備及條件 1.電動機(jī)兩臺； 2.自動空氣開關(guān)、交流接觸器、熱繼電器、互感器、按鈕等常用電器； 3.FX2-64MR可編程控制器； 4.A/D模塊、A/D模塊； 5.萬用表、電烙鐵等常用工具； 6.計(jì)算機(jī)一臺。 任務(wù)下達(dá)時間： 2015年12月28日 畢業(yè)設(shè)計(jì)開始與完成時間： 2015年12月28日至 2016年05 月22日 組織實(shí)施單位： 教研室主任意見： 簽字： 2015年12月30日 院領(lǐng)導(dǎo)小組意見： 簽字： 2015 年12月31日 633 High Performance Induction Motor Control Via Feedback Linearization M. P. Kazmierkowski and D. L. Sobczuk Institute of Control and Industrial Electronics, Warsaw University of Technology, u l . Koszykowa 75, 00-662 WarszawaPoland Phone: +48/2/6280665; Fax: +48/2/6256633 E-mail: mpkov.isep.pw.edu.pl; sobczukov.isep.pw.edu.pl Abstract - This paper presents a feedback linearization approach for high performance induction motor control. The principle of the method is discussed and compared with most popular in AC motor drive technology field oriented control technique. Some oscillograms illustrating the properties of the PWM inverter-fed induction motor with control via feedback linearization are presented. INTRODUCTION The induction motor thanks to its well known advantages as simply construction, reliability, raggedness and low cost has found very wide industrial applications. Furthermore, in contrast to the commutator dc motor, it can also be used in aggressive or volatile environments since there is no prob- lems with spark and corrosion. These advantages, however, are occupied by control problems when using induction motor in speed regulated industrial drives. Ths is due primarily three reasons: (a) - the induction motor is high order nonlinear dynamic system with internal coupling, (b) - some state variables, rotor currents and fluxes, are directly not measurable, (c) - rotor resistance (due to heating) and magnetising inductance (due to saturation) varies consider- ably with a sigluficant impact on the system dynamics. The most popular high performance induction motor control method known as Field Oriented Control (FOC) or Vector Control has been proposed by Hasse 4 and Blaschke 11. In this method the motor equation are (rewritten) transformed in a coordmate system that rotates with the rotor flux vector. These new coordinates are calledfield coordinates. In field coordinates - for the constant rotor flux amplitude - there is a linear relationshp between control variables and speed. Moreover, as in separately excited DC motor, the reference for the flux amplitude can be reduced in field weakening region in order to limit the stator voltage at high speed. Transformation of the induction motor equations in the field coordinates has a good physical basis because it corresponds to the decoupled torque production in separately excited DC motor. However, from the theoretical point of view other type of coordinates can be selected to achieve decoupling and linearization of the induction motor equations. Krzeniinski 7 has proposed a nonlinear controller based on multiscalar motor model. In this approach, similarly as in field oriented controller, it is assumed that the rotor flux IEEE Catalog Number: 95TH8081 amplitude is regulated to a constant value. Thus, the motor speed is only asymptotically decoupled from the rotor flux Bodson et al. 2,3 have developed a nonlinear control sys- tem based on iiiput-output linearization. In this system, the motor speed and rotor flux are decoupled exactly. The system, however, use the transformation in field coordinates. Marino et al. 8,9 have proposed a nonlinear transformation of the motor state variables, so that in the new coordinates, the speed and rotor flux amplitude are decoupled by feedback. Similar transformation have been used by Sabanovic et al. E for decoupled rotor flux and speed sliQng mode controller. In the paper the feedback linearization control of induction motor is presented. In contrast to the works 2,3 the block diagrams and relationships to field oriented control are dis- cussed. Also, figures illustrated the properties of the control system when the motor is fed by PWM inverter are shown. MATHEMATICAL MODEL OF THE INDUCTION MOTOR Mathematical description of the induction motor is based on complex space vectors, which are defined in a coordinate system rotating with angular speed oK. In per unit and real- time representation the following vectorial equations describe behaviour of the motor 6: The electromagnetic torque m can be expressed as In the case of the squirrel-cage induction motor, the rotor voltage-vector vanishes from Eq. 2, having zero value. If a 633 current controlled PWM inverter is used, the stator voltage Eq. 1 can be neglected because it does not affect the control dynamics of the drive. FIELD ORIENTED CONTROL (FOC) In the case of field oriented control, it is very convenient to select the angular speed of the coordmate system oK equal a , . Under these assumptions, substituting the rotor current vector from the rotor voltage Eq. 2 by Eq. 4 ve obtain a differential equation for the rotor flux vector: Equations Eq. 10, Eq. 12 and Eq. 5 form the block diagram of the induction motor in the field oriented coordmates x y (Fig. 1). Diagram of control system applied to induction motor (direct field orientation) is.presented in Fig 2. In many cases as flux, speed, i , , i , controllers, simple PI regulators are used. FEEDBACK LTNEARTZATION CONTROL (FLC) Using p.u. time we can write the induction motor equations in the following form 6,11: (7)- where T , is the rotor time constant expressed as x =f(x + %.ga + usp g p (13) X Tr=ITN rr For the field oriented coordinates x-y we have Wm= Yr y = o ry where and m. I Note that am, yra, yq are not dependent on control signals U , , U$. In this case it is easily to choose two variables dependent on x only. For example we can define 5,10. Fig 1 Block diagram of induction motor in x-y field coordinates Eq. 10 describes the influence of the flux stator current according to Eq 5, can be expressed as follows: components i, on the rotor flux. The motor torque, XI = Vra2 + vr3 = V: (18) &(x) = 0, (1 9) 635 d P-P 1 sa 1 SG - 1rp Vector 1 Sigoals . +r ffiatioc &a Estimaiion - Transfor- vr* * vrB * % Fig. 2. Control of induction motor via field orientation Let +,(x), I$(X) are the output variables. The aim of control is to obtain: 0 constant flux amplitude, 0 reference angular speed. P a r t of the new state variables we can choose according to Eq. IS, Eq. 19. So the full definition of new coordinates are given by 8,9: In the further part of this section w e will consider the system consists of the first fourth equations. Note, that the fifth equation is as follows: We can rewite the remain system Eq. 22 in the following form: D is given by: After simple calculations one can obtain: 6 where D-I we can calculate using the following formula It is easily to show that if+l f 0 then det(D) # 0. In this case we can define linearizing feedback as: The resulting system is described by the equations: 2, =z2 z,= , z3 = z4 2 , = v 2 Block diagram of induction motor with new control signals is presented in Fig 3. Control signals v1 ,v2 one can calculate using linear feedback: where coefficients k, k, k, k, are chosen to determinate closed loop system dynamic. Control algorithm consists of two steps: 0 calculations v1 ,v2 according to Eq. 32, Eq 33, 0 calculations U , , uSp according to Eq. 29. Diagram of control system applied to induction iiiotor (feedback linearization) is presented in Fig. 4. Fig. 4. Control of induction motor via feedback linearization 637 -0461 1 -09El RESULTS -04FI , -09.! The simulated oscillogranis obtained for FLC and FOC sys- tems with linear speed and rotor flux controllers (motor, in- verter and controllers data are given in Appendix) are shown in Fig. 5. These oscilllograms, show the speed reversal over the constant flux amplitude and field weakening ranges when motor is fed from VSI inverter with sinusoidal PWM. As can be seen from Fig. 5B the field oriented control does not guarantee full decoupling betwen speed and flux of the motor. With linear speed controller the FOC systein imple- ments torque current limitation, whereas the FLC system limits the motor torque (see Fig. 5A). Therefore, in the FOC system the torque is reduced in field weakening region and the speed transient is slower in FLC system. A-FLC B - FOC lineat r O C 6 1 C 2 0.3 6 4 05,-0.0 012 0.24 0.36 048 0 8 Cl ; 0 10 05 05 00 00 -0 5 -0 5 - 1 c -! 0 d)oC 0 1 0 2 03 0 4 0 5 d)OO 012 024 036 018 06 j -0961 I c 9f 1 t)C7 L 1 ( 7 0 3 0 4 5 e ) O 0 Ci 024 036 048 O t O 01 D: 03 0 4 05 0 0 O l i 024 036 0 4 B 06 Flg 5 Conk01 of mduclon motor via feedback linearization and field oriented control (Speed reversal mcludmg field weakening range). a) actual and reference speed (comer con,) b) torque m. c) flux component and amplrtude ( y,. yr), d) flux current isX. e) torque current i s , , . f ) current component lso To guarantee full decoupling in FOC system working with field weakening region a PI speed controller with nonlinear part (controller output signal should be divided over the rotor flux amplitude mrdyr) has to be applied. This division compensates for the internal multiplication (m = vr i ) sv needed for motor torque production in field oriented coordi- nates (Fig. 1 .). With such a nonlinear speed controller a very similar behaviour to FLC can be achieved (see Fig. 6.). In Fig. 7. the response to speed reference change for constant flux amplitude is presented. Note, that in contrast to the FOC system (where the control variables are U , , u ) , the control variables in FLC system (vl, VJ are exactly decoupled Fig. 6. Control of mdudion motor ia feedback linearization and field oriented control with nonlniear term (Speed reversal including field weak&y range): a) adual and reference speed (omrep om) b) torque m, c) flux component and amplitude (iy,. t y , ) . d) flux current isx. e) torque current is,., 9 cunent component i SP A - FOC 0) ;y 04 t,CO 003 ODE 009 12 GI5 -, 1 2 0 6 d)G0 003 006 005 G I ? 0 1 5 ) G O 003 GO6 00s 012 015 0 6 00 -0 6 -0 E - 1 i -1 2 e)00 003 GO& 009 C l I Ol5e)GC 003 GO6 005 G12 0 0 0 i I 0 0 003 006 003 012 D!: C C 063 031 005 01; 0 : Fig 7. Control of mdudion motor via feedback lmeariiatton and field onitcd conk01 (the response to speed rcfercnce dmigc - mstmt f l u s raigc) a) adual and refaence speed (o um). b) refercnce control agial A - u s , . B - vl. c) reference mtrof%&l A - u s ) . . B - v2. d) flux currail ish e) torque currat I 6) 638 CONCLUSIONS Kurzchlusslauferniotoren, Reglungstechnik, 20: pp. 60- 66. 1972. In this work a high performance Feedback Linearization Control (FLC) system for PWM inverter-fed induction motor drives is presented. The block diagrams and relationshps to conventional Field Oriented Control CFOC) are discussed. SI A Isidori: Nonlinear Control Systems, Comniunica- tions and Control Engineering. Springer Verlag, Berlin, second ehtion, 1989. .I The main features and advantages of the presented control 6 M. P. Kaimierkomki and H. Tunia: Automatic Control systems can be summarised as follows: of Converter-Fed Drives, ELSEVIER Amsterdam- with control variables v, 17, the FLC guarantee the exactly London-New York-Tokyo, 1994. decoupling of the motor speed and rotor flux control in both dynamic and steady states. Therefore, lugh performance drive system working in both constant and field weakening range can be implemented using a linear speed and flux 7 Z. Krzeminski: Multi-scalar models of an induction motor for control system synthesis, Scieiiia Electrica, 33(3): pp. 9-22, 1987. controllers. Adaptive partial with control variables is, is, the FOC cannot guarantee feedback linearization of induction motors, In Pro- the exactly decoupling of the motor speed and rotor flux ceedings o f the 29th Conference on Decision aid control in dynamic states. Therefore, high performance drive Control, Honolulu, Hawaii, pp. 3313-3318, Dec. 1990. 8 R. Marino, S. Peresada, and P. Valigi: 9 R. Marino and P. Valigi: Nonlinear control of induction motors: a simulation study, In European Coiitrol Corifereme. Greiioble, France, pp. 1057-1062, 1991, system working in both constant and field wakening range requires a speed controller with nonlinear (division over the rotor flux amplitude) part. FLC is implemented in a state feedback fashion and needs more complex signal processing (full information about motor state variables and load torque is required). Also, the lo H. Nijmeijer and A van der Schaft: Nonlinear dy- namical control systems, Springer Verlag, 1990. ll D. L. Sobczuk: If Nonlinear control for induction nie transformation and new control variables vl, v2 used in FLC have no so dlrect physical meaning as in, is, (flux and torque current, respectively) in the case of FOC system. tor, In Proceedings PEIIK 94, pp. 684-689, 1994. FOC can be implemented in classical cascade control (121 A Sabanovic and D. B. Izosiniov: Applications of structure and, therefore, an overload protection can easy be sliding modes to induction motor control, IEEE achieved using reference currents limiters on the outputs of Transaction on Industry Applications, 17( 1): pp. 41-49, 1981. It can be expected, however, that - because in FLC vari- ables (am, v:) and its derivative (Om, W,) are used as new coordinates - this approach will .be well suited for sliding mode speed and position controllers. Therefore, FLC create an interesting alternative to FOC for applications where high performance induction motor drive system are required. the flux and speed controllers, respectively. APPENDIX Motor Data: Controllers Data: REFERENCES rr = 0.0464 p.u. rs = 0.0314 p.u. Feedback linearization: k, = 2.25 p.u. 11 F. B1aschke:Da.s Verfahren der Feldorientirung zur xr = 2.237 PU. xs = 2.194 P.U. XM = 2.133 P.U. TM= 0.2 s k, = 3.0 p.u. k21 = 6.25 p.u. kZz = 5.0 p.u. Reglung der Asynchronmaschne, Siemens Forschungs- . und Entwicklungsber, l(1): pp. 184-193, 1972 2 M. Bodson, J. Chiasson, and R. Novotnak: High per- - _ _ formance induction motor control via input-output line- arization, IEEE Control Systems, pp. 25-33, August 1994. Data: Field Orientation: Flux PI Controller: d = 4.0 P.U. * KR= 10; TR=318 ms Tv-50ps Speed PI controller: 3 .J. Chiasson, A Chaudhari, and M. Bodson: Nonlinear controllers for the induction motor, In IFAC Nonlinear Control System Design Symposium. Bordoeaux France, TR=318 ms pp. 150-155, 1992. Current controllers: K, = 150; 4 K. Hasse: Drehzahlgelverfahren fur schnelle Um- kehrantriebe mit stromrichtergespeisten Asynchron -